Phase locked loop circuit with oscillator signal based on switched impedance network

ABSTRACT

Phase-locked loop circuitry to generate an output signal, the phase-locked loop circuitry comprising oscillator circuitry, switched resistor loop filter, coupled to the input of the oscillator circuitry (which, in one embodiment, includes a voltage-controlled oscillator), including a switched resistor network including at least one resistor and at least one capacitor, wherein an effective resistance of the switched resistor network is responsive to and increases as a function of one or more pulsing properties of a control signal (wherein pulse width and frequency (or period) are pulsing properties of the control signal), phase detector circuitry, having an output which is coupled to the switched resistor loop filter, to generate the control signal (which may be periodic or non-periodic). The phase-locked loop circuitry may also include frequency detection circuitry to provide a lock condition of the phase-locked loop circuitry.

RELATED APPLICATIONS

This non-provisional application is a divisional of U.S. patentapplication Ser. No. 16/505,469, filed Jul. 8, 2019, entitled“Integrated Circuit With Oscillator Signal Based On Switched-ResistanceCircuitry,” which is a divisional of U.S. patent application Ser. No.16/055,934, filed Aug. 6, 2018, entitled “Phase Locked Loop WithSwitched-Component Loop Filter” (now U.S. patent Ser. No. 10/389,371),which is a divisional of U.S. patent application Ser. No. 15/284,302,filed Oct. 3, 2016, entitled “PLL With Accelerated Frequency Lock” (nowU.S. Pat. No. 1,063,244) which is a divisional of U.S. patentapplication Ser. No. 14/938,528, filed Nov. 11, 2015, entitled“High-Gain Locked-Loop Phase Detector” (now U.S. Pat. No. 9,461,658),which is a divisional of U.S. patent application Ser. No. 14/466,988,filed Aug. 23, 2014, entitled “Phase Locked Loop Circuitry HavingSwitched Resistor Loop Filter Circuitry, and Methods of Operating Same”(now U.S. Pat. No. 9,203,417), which is a divisional of U.S. patentapplication Ser. No. 13/653,640, filed Oct. 17, 2012, entitled “PhaseLocked Loop Circuitry Having Switched Resistor Loop Filter Circuitry,and Methods of Operating Same” (now U.S. Pat. No. 8,829,959), which is adivisional of U.S. patent application Ser. No. 12/860,875, filed on Aug.21, 2010, entitled “Phase Locked Loop Circuitry Having Switched ResistorLoop Filter Circuitry, and Methods of Operating Same” (now U.S. Pat. No.8,299,826), which claims the benefit of U.S. Provisional Application No.61/236,682, entitled “Switched Resistor Loop Filter Circuitry, LowArea/Power Implementation of Phase Locked Loop Circuits, and Methods ofManufacturing and Operating Same,” filed Aug. 25, 2009. Each of theforegoing patent applications is hereby incorporated by reference hereinin its entirety.

INTRODUCTION

Phase-locked loop (PLL) circuits are a key component of most moderncommunication circuits, and are also used in a variety of digitalprocessor applications in order to generate high frequency, low jitterclock sources. While there are a wide variety of PLL architectures thathave been proposed, the charge pump PLL, which is shown in FIG. 1A, hasemerged as the most popular analog implementation, and is composed of aphase/frequency detector (PFD), charge pump, RC network, voltagecontrolled oscillator (VCO), and frequency divider. More recently, thedigital PLL structure, shown in FIG. 1B, has gained attention due to itshigh compatibility with advanced CMOS processes, and is composed of atime-to-digital converter (TDC), digital loop filter, digitallycontrolled oscillator (DCO), and frequency divider.

While the existing analog and digital PLL structures are now well provenand can satisfy the performance needs for a wide variety ofapplications, they have shortcomings when trying to achieve completeintegration of the PLL with low silicon area and low power consumption.On the analog side, charge pump PLLs are well known to require a verylarge capacitor in their loop filter, which directly impedes the goal ofachieving low silicon area. In fact, it is not uncommon for thecapacitor area requirements to be so large that it must be placedoff-chip, which negatively impacts the noise sensitivity of the PLL andadds to manufacturing cost of the intended product for the PLL. Inaddition, 1/f noise from the charge pump is often a significant issuewhen seeking low phase noise for the PLL. On the digital side, anadvantage of the digital PLL is that its loop filter can be quite smallin area and easily configured for a wide variety of applications, and awell designed TDC can achieve low 1/f noise. However, since a digitalPLL contains many signals that continuously switch between supply andground, achieving very low power consumption can be challenging,especially if older CMOS fabrication processes, such as 0.18 u CMOS, areutilized.

When striving for both low power and low area, one could first decidewhether the analog or digital strategy provides the best starting point.A key insight is that simplicity in design, i.e., minimizing the numberof “moving parts”, will typically benefit both of these goals. As such,we propose advancing the analog approach by seeking dramatic reductionof the loop filter capacitor and reduced impact of 1/f noise. Inparticular, while not exhaustive of the inventions described and/orillustrated herein, we will discuss and/or illustrate new techniques andcircuitry in the areas of:

-   -   1) switched resistor networks, architectures, topologies and        circuitry to, for example, implement in a loop filter,    -   2) switched resistor loop filter architectures, topologies and        circuitry,    -   3) phase detector architectures, topologies and circuitry that        reduce the impact of loop filter noise by achieving high gain in        their detection characteristic, and    -   4) methods of frequency detection—for example, frequency        detection methods that “leverage” switched capacitors and        digital logic to achieve rapid frequency acquisition in robust        manner.

SUMMARY

There are many inventions described and illustrated herein. The presentinventions are neither limited to any single aspect nor embodimentthereof, nor to any combinations and/or permutations of such aspectsand/or embodiments. Moreover, each of the aspects of the presentinventions, and/or embodiments thereof, may be employed alone or incombination with one or more of the other aspects of the presentinventions and/or embodiments thereof. For the sake of brevity, many ofthose permutations and combinations will not be discussed separatelyherein.

Importantly, the present inventions are neither limited to any singleaspect nor embodiment, nor to any combinations and/or permutations ofsuch aspects and/or embodiments. Moreover, each of the aspects of thepresent inventions, and/or embodiments thereof, may be employed alone orin combination with one or more of the other aspects and/or embodimentsthereof. For the sake of brevity, certain permutations and combinationsare not discussed and/or illustrated separately herein.

In a first principle aspect, certain of the present inventions aredirected to phase-locked loop circuitry to generate an output signal,the phase-locked loop circuitry comprising oscillator circuitry havingan input and an output, switched resistor loop filter, having an outputwhich is coupled to the input of the oscillator circuitry (which, in oneembodiment, includes a voltage-controlled oscillator), the switchedresistor loop filter includes a switched resistor network including atleast one resistor and at least one capacitor, wherein an effectiveresistance of the switched resistor network is responsive to andincreases as a function of one or more pulsing properties of a controlsignal (wherein pulse width and frequency (or period) are pulsingproperties of the control signal). The phase-locked loop circuitry ofthis aspect of the inventions further includes phase detector circuitry,having an output which is coupled to the switched resistor loop filter,to generate the control signal (which may be a periodic or non-periodicsignal).

In one embodiment, one of the pulsing properties of the control signalis a pulse on-time and the pulse on-time of the control signal isshorter than an RC time constant of the switched resistor network. Inanother embodiment, the effective resistance of the switched resistornetwork is determined by an average pulse width and average frequency ofthe control signal. Here, the control signal may be non-periodic.

The phase-locked loop circuitry of this aspect of the inventions mayalso include divider circuitry, disposed in a feedback path, to couplethe output of the oscillator circuitry to an input of the phase detectorcircuitry.

In another aspect, the present inventions are directed to phase-lockedloop circuitry to generate an output signal, the phase-locked loopcircuitry comprising oscillator circuitry (which, in one embodiment,includes a voltage-controlled oscillator), switched resistor loopfilter, coupled to the input of the oscillator circuitry, the switchedresistor loop filter includes a switched resistor network including aplurality of resistors, a plurality of capacitors and at least oneswitch, wherein an effective resistance of the switched resistor networkis responsive to and increases as a function of one or more pulsingproperties of a control signal, and phase detector circuitry, having anoutput which is coupled to the switched resistor loop filter, togenerate the control signal (which may be a periodic or non-periodicsignal).

In this aspect of the present inventions, the phase-locked loopcircuitry also includes divider circuitry. In one embodiment, thedivider circuitry includes a division factor so that the phase-lockedloop circuitry, in operation, provides an integer-N multiple of thereference frequency, where N is greater than 1. In another embodiment,the divider circuitry includes a division factor so that thephase-locked loop circuitry, in operation, provides a fractional-Nmultiple of the reference frequency, where the integer portion, N, isgreater than 0 and the fractional component is within the range of 0 to1.

In one embodiment, one of the pulsing properties of the control signalis a pulse on-time and the pulse on-time of the control signal isshorter than an RC time constant of the switched resistor network. Inanother embodiment, the effective resistance of the switched resistornetwork is determined by an average pulse width and average frequency ofthe control signal.

Notably, the oscillator circuitry, switched resistor loop filter, phasedetector circuitry and divider circuitry may be portions of a monolithicintegrated circuit device.

In yet another aspect, the present inventions are directed tophase-locked loop circuitry comprising oscillator circuitry, switchedresistor loop filter, coupled to an input of the oscillator circuitry,the switched resistor loop filter includes a switched resistor networkincluding a plurality of resistors, a plurality of capacitors and atleast one switch, wherein an effective resistance of the switchedresistor network is responsive to and increases as a function of one ormore pulsing properties of a control signal, phase detector circuitry,having an output which is coupled to the switched resistor loop filter,to generate the control signal (which may be periodic or non-periodic)and divider circuitry to couple the output of the oscillator circuitryto an input of the phase detector circuitry.

In this aspect, the phase-locked loop circuitry also includes frequencydetection circuitry, coupled to the output of the divider circuitry, toprovide a lock condition of the phase-locked loop circuitry. Thefrequency detection circuitry includes (i) circuitry to generate asignal which is representative of the frequency of the output signal ofthe phase-locked loop circuitry, (ii) comparison circuitry to comparethe signal which is representative of the frequency of the output signalof the phase-locked loop circuitry to a reference input to thephase-locked loop circuitry, and (iii) a switched capacitor networkincluding at least one capacitor.

In one embodiment, the circuitry of the frequency detection circuitryincludes a counter to generate the signal which is representative of thefrequency of the output signal of the phase-locked loop circuitry.

In another embodiment, one of the pulsing properties of the controlsignal is a pulse on-time such that the pulse on-time of the controlsignal is shorter than an RC time constant of the switched resistornetwork.

In yet another embodiment, the effective resistance of the switchedresistor network is determined by an average pulse width and averagefrequency of the control signal. In this embodiment, the control signalmay be non-periodic.

As stated herein, there are many inventions, and aspects of theinventions, described and illustrated herein. This Summary is notexhaustive of the scope of the present inventions. Indeed, this Summarymay not be reflective of or correlate to the inventions protected by theclaims in this or continuation/divisional applications hereof.

Moreover, this Summary is not intended to be limiting of the inventionsor the claims (whether the currently presented claims or claims of adivisional/continuation application) and should not be interpreted inthat manner. While certain embodiments have been described and/oroutlined in this Summary, it should be understood that the presentinventions are not limited to such embodiments, description and/oroutline, nor are the claims limited in such a manner (which should alsonot be interpreted as being limited by this Summary).

Indeed, many other aspects, inventions and embodiments, which may bedifferent from and/or similar to, the aspects, inventions andembodiments presented in this Summary, will be apparent from thedescription, illustrations and claims, which follow. In addition,although various features, attributes and advantages have been describedin this Summary and/or are apparent in light thereof, it should beunderstood that such features, attributes and advantages are notrequired whether in one, some or all of the embodiments of the presentinventions and, indeed, need not be present in any of the embodiments ofthe present inventions.

BRIEF DESCRIPTION OF THE DRAWINGS

In the course of the detailed description to follow, reference will bemade to the attached drawings. These drawings show different aspects ofthe present inventions and, where appropriate, reference numeralsillustrating like structures, components, materials and/or elements indifferent figures are labeled similarly. It is understood that variouscombinations of the structures, components, materials and/or elements,other than those specifically shown, are contemplated and are within thescope of the present inventions.

Moreover, there are many inventions described and illustrated herein.The present inventions are neither limited to any single aspect norembodiment thereof, nor to any combinations and/or permutations of suchaspects and/or embodiments. Moreover, each of the aspects of the presentinventions, and/or embodiments thereof, may be employed alone or incombination with one or more of the other aspects of the presentinventions and/or embodiments thereof. For the sake of brevity, many ofthose permutations and combinations will not be discussed and/orillustrated separately herein.

FIG. 1A is a schematic block diagram representation of a classicalcharge pump PLL;

FIG. 1B is a schematic block diagram representation of a digital PLL;

FIG. 2A illustrates a block diagram representation of an exemplaryembodiment of switched resistor phase-locked loop (PLL)structure/circuitry according to certain aspects of the presentinventions;

FIG. 2B illustrates a modeling of an exemplary switched resistorphase-locked loop (PLL) structure/circuitry according to certain aspectsof the present inventions;

FIGS. 3A-3C illustrate the relationship of 1/f noise to variousconfigurations wherein FIG. 3A corresponds to high 1/f noise from MOScurrent mirror yields high noise in charge pump, FIG. 3B corresponds toreduction of 1/f noise in charge pump by using resistor degeneration,FIG. 3C corresponds to dramatic reduction of 1/f noise by using proposed“switched resistor” technique instead a MOS current mirror;

FIGS. 4A-4C depict the impact of switching characteristics on effectiveresistor value, wherein FIG. 4A corresponds to a pulse waveform forturning on and off resistor switch, FIG. 4B (ideal switched resistor)corresponds to ideal multiplication effect of switching waveform, andFIG. 4C (practical switched resistor) corresponds to reducedmultiplication effect of switching waveform due to parasiticcapacitance;

FIGS. 5A and 5B illustrate exemplary embodiments of loop filtersaccording to certain aspects of the present inventions wherein theexemplary loop filters employ switched resistor embodiments according tocertain aspects of the present inventions; notably, FIG. 5A is atopology that includes high comparison frequency and wide PLL bandwidth(High Reference Frequency), and FIG. 5B is a topology that includes amulti-phase approach providing high rejection of reference spurs (LowReference Spur);

FIGS. 6A and 6B illustrate modeling of phase detector gain and overallPLL dynamics, wherein FIG. 6A is associated with a classical charge pumpPLL, and FIG. 6B is associated with a switched resistor PLL;

FIG. 7 illustrates a conventional tristate PFD and corresponding signalsand phase detector characteristic;

FIGS. 8A and 8B illustrate exemplary embodiments of phase detectorcircuitry according to certain aspects of the present inventionsincluding, for example, high gain phase detectors for the switchedresistor loop filters of FIGS. 5A and 5B notably, FIG. 8A may becharacterized as a high speed phase detector which may be implemented inconjunction with, for example, the loop filter FIG. 5A, and FIG. 8B maybe characterized as a multi-phase phase detector which may beimplemented in conjunction with, for example, the loop filter FIG. 5B;

FIG. 9 illustrates an exemplary embodiment of asynchronous dividercircuit and a technique of generating short output pulses therefrom,wherein the output pulses include a variable width;

FIG. 10 illustrates a block diagram and gate and CMOS transistor leveldivide-by-2/3 stage asynchronous divider circuit;

FIG. 11 illustrates the timing relationship of an asynchronous dividerwith four times the frequency of the reference for multi-phase pulsegeneration;

FIG. 12 illustrates an exemplary divider circuit and a method ofcontrolling the four divide values to achieve a divide value of N;

FIG. 13 illustrates an exemplary embodiment of multi-pulse phasedetector circuitry to generate signals N₀ through N₃, notably, anadvantage of this approach is that the divide values can be aligned asdesired to their respective place within the reference period;

FIG. 14 illustrates an exemplary embodiment of circuitry includingbuffer delays to provide desired pulse widths and separation of theoutput signals, along with frequency division to control the period of aswitching signal for a given resistor; notably, buffer delays 1, 2, and3 are used to create separation between different pulse signals in orderto provide non-overlapping behavior between the on-times of thosepulses, whereas buffer delay 4 is employed to control the width of thepulses in the Last signal, and a divide-by-N circuit is used to set theperiod of those pulses in increments of the period of Up/Down pulses;

FIG. 15 illustrates an exemplary embodiment of circuitry includingbuffer delays to provide exemplary pulse widths and separation of theoutput signals, along with frequency division to control the period of aswitching signal for a given resistor, notably, this circuitry generatesmulti-phase pulses in which the width of Last pulse corresponds to thepulse width of the divider output;

FIG. 16 illustrates an exemplary embodiment of frequency detectioncircuitry wherein a range of variation on the output voltage of the loopfilter as the phase detector characteristic is swept through due to afrequency offset; notably, this circuit highlights the issues of using aswitched resistor network in which high frequency attenuation occursfrom the phase detector output to VCO tuning input;

FIG. 17 illustrates an exemplary embodiment of frequency detectioncircuitry wherein a range of variation on the output voltage of the loopfilter as the phase detector characteristic is swept through due to afrequency offset; notably, this exemplary frequency detection circuitand method employs edge count comparison between reference and divideroutput and a switched capacitor network to adjust loop filter output;moreover, the inventive circuitry and methods of frequency detectionwhich employ a switched capacitor network as described and illustratedherein may be implemented in conjunction with a wide range of phasedetectors and loop filters, including a switched resistor loop filterand high gain phase detectors as described and illustrated herein andconventional PLL loop filters and phase detectors including, forexample, charge pump PLL loop filters and phase detectors;

FIG. 18 illustrates an exemplary embodiment of frequency detectorcircuitry wherein the circuitry compares the edge locations of thereference as it propagates through cascaded registers which are clockedby the divider output; notably, this frequency detector circuitry may bewell suited for the multi-pulse phase detector shown in FIG. 8B; itshould also be noted that the frequency detection approach suggested inthis figure may be implemented in conjunction with phase detectors otherthan those discussed and illustrated herein;

FIG. 19 illustrates a system level simulation of frequency acquisitionof a switched resistor PLL using the frequency detector of FIG. 18;

FIG. 20 illustrates good versus bad phase alignment between edges whencounting edges for frequency detection;

FIG. 21 illustrates an exemplary embodiment of phase detector circuitry(which is modified relative to the phase detector circuitry of FIG. 8A)which facilitates robust frequency detector edge counting;

FIG. 22 illustrates the frequency detection logic/circuitry of FIG. 18in combination with the phase detector and switched resistor loop filtercircuitry of FIG. 21;

FIG. 23 illustrates a linearized model of a switched resistor PLL with aloop filter of FIG. 5B and a multi-pulse phase detector of FIG. 8B;

FIG. 24 is a representation of modeling of noise in a switched resistorloop filter;

FIG. 25 is a block diagram representation of a noise analysis for aswitched resistor PLL;

FIG. 26 illustrates instantaneous voltage waveform of RC network asimpacted by switched resistor operation;

FIG. 27 illustrates exemplary noise folding of quantization noise fromdivider dithering due to nonlinearity in a switched resistorfractional-N PLL

FIG. 28 illustrates an exemplary embodiment of switched resistor PLLtopology which lowers the impact of nonlinearity while still allowingwide PLL bandwidth;

FIG. 29 illustrates an exemplary embodiment of configurable switchedresistor loop filter to allow for a range of PLL bandwidths;

FIG. 30 illustrates exemplary varactor range considerations for aswitched resistor PLL;

FIG. 31 illustrates certain design consideration/issues for a switchedresistor PLL implementation;

FIG. 32 illustrates phase noise plot from running Matlab script inAppendix 1 of the Provisional Application;

FIGS. 33A and 33B illustrate block diagram representations of exemplaryembodiments of switched resistor phase-locked loop (PLL)structure/circuitry wherein control signal generator circuitry generatescontrol or switching signals to control the effective resistance of theswitched resistor network and/or the switched resistor loop filter,according to certain aspects of the present inventions; notably, thesignal generator circuitry may be integrated in the phase detectorcircuitry (FIG. 33A) or separate from the phase detector circuitry (FIG.33B).

Again, there are many inventions described and illustrated herein. Thepresent inventions are neither limited to any single aspect norembodiment thereof, nor to any combinations and/or permutations of suchaspects and/or embodiments. Each of the aspects of the presentinventions, and/or embodiments thereof, may be employed alone or incombination with one or more of the other aspects of the presentinventions and/or embodiments thereof. For the sake of brevity, many ofthose combinations and permutations are not discussed separately herein.

DETAILED DESCRIPTION

At the outset, it should be noted that there are many inventionsdescribed and illustrated herein. The present inventions are neitherlimited to any single aspect nor embodiment thereof, nor to anycombinations and/or permutations of such aspects and/or embodiments.Moreover, each of the aspects of the present inventions, and/orembodiments thereof, may be employed alone or in combination with one ormore of the other aspects of the present inventions and/or embodimentsthereof. For the sake of brevity, many of those permutations andcombinations will not be discussed separately herein.

Further, in the course of describing and illustrating the presentinventions, various structures, components, materials and/or elements,as well as combinations and/or permutations thereof, are set forth. Itshould be understood that structures, components, materials and/orelements other than those specifically described and illustrated, arecontemplated and are within the scope of the present inventions, as wellas combinations and/or permutations thereof.

Notably, certain aspects and/or embodiments of the present inventions(including certain components, materials and/or elements of the presentinventions, as well as combinations and/or permutations thereof) aredescribed and illustrated in EXHIBITS 1 and 2 of the ProvisionalApplication. Such aspects and/or embodiments may be in lieu of or inaddition to those described and illustrated herein. Indeed, the aspectsand/or embodiments of the present inventions described and illustratedin EXHIBITS 1 and 2 of the Provisional Application may provide furthertechnical support for those aspects and/or embodiments of the presentinventions described and illustrated herein. EXHIBITS 1 and 2 of theProvisional Application are to be considered as a portion of thedescription of the present inventions.

With that in mind, in one aspect, the present inventions are directed toa switched resistor network for phase-locked loop (PLL)structure/circuitry. In another aspect, the present inventions aredirected to a switched resistor phase-locked loop (PLL)structure/circuitry. (See, for example, FIGS. 2A and 2B). In yet anotheraspect, the present inventions are directed to techniques to implement alow area and/or low power switched resistor PLL structure/circuitry. Thepresent inventions are also directed to methods of operating,controlling and manufacturing PLL circuitry according to one or moreaspects and/or embodiments of the present inventions.

In another aspect, the present inventions are directed to switchedcapacitor frequency detection circuitry and techniques. When implementedin conjunction with PLL circuitry (for example, whether the PLLcircuitry of the present inventions or conventional type PLL circuitry,for example, circuitry employing a charge pump circuit), the frequencydetection circuitry provides a rapid and robust lock condition of thePLL circuitry.

Notably, PLL structures/circuitry according to one or more aspectsand/or embodiments of the present inventions may include directconnection of a passive RC network to a phase detector (PD) without acharge pump to, for example, improve 1/f noise performance, pulsing ofone or more resistor elements to increase the associated resistancevalue, high gain PD implementations to compensate for the low DC gain ofthe passive loop filter and to improve PLL noise performance, and anefficient frequency detection method using a switched capacitor networkto achieve relatively robust and rapid frequency acquisition.

A. Exemplary Switched Resistor Network

The switched resistor network, according to at least certain embodimentsof the present inventions, employs switches (for example, CMOS devices)to gate current into a resistor network (for example, one or moreresistor elements). A benefit of doing so is to lower the impact of 1/fnoise, as indicated by FIGS. 3A-3C. As shown in FIG. 3A, a basic chargepump consists of a current mirror whose output is switched on or off.Unfortunately, as mentioned earlier, the CMOS devices have substantial1/f noise which will directly degrade the low frequency noiseperformance of the charge pump and the overall PLL. As shown in FIG. 3B,we can reduce the impact of the CMOS 1/f noise by degenerating the CMOSdevices with resistor elements such as polysilicon resistors. Since suchresistors have negligible 1/f noise in comparison to CMOS devices, the1/f noise can be reduced in proportion to the amount of voltage that isdropped across the resistor (i.e., higher voltage drops yields lower 1/fnoise). As shown in FIG. 3C, a version of resistor degeneration dropsall of the voltage across the resistor by eliminating the CMOS currentmirror device. Ideally, 1/f noise is eliminated in this embodiment, butin practice the switch, which typically corresponds to a CMOS device inits triode region, will have a finite voltage drop across it, andtherefore will add some 1/f noise. Also, in the embodiment where theresistor is connected to a power supply node, the 1/f noise of the powersupply will also have an impact on the current through the resistor.However, in either of these embodiments, the degeneration offered by theresistor should significantly reduce the impact of such 1/f noisecompared to classical charge pump structures.

The effective resistance of switching a resistor may be characterized asa function of the pulse duration and frequency of the switching action.An intuitive explanation of this behavior is that given a specifiedvoltage across the resistor, the average current through the resistorwill be reduced as the pulse width and/or pulse frequency is reduced,which directly increases its effective resistance. FIG. 4A illustratesan exemplary parameterized view of a pulse waveform driving the switchon and off, and FIG. 4B illustrates the effective resistance that wouldideally be achieved as a function of the pulse width (T_(on)) andfrequency (1/T_(period)). Indeed, the pulse width and frequency (orperiod) are “pulsing properties” of the switching signal. Note that verylarge effective resistance values can be achieved in theory, and acontinuous range of values supported by appropriate choice of T_(on) andT_(period). Notably, the functional relationship of the effectiveresistance of switching a resistor to the pulse duration and frequencyof the switching action may be linear or non-linear.

While FIG. 4A illustrates or suggests the pulse waveform as a periodicsignal, the signal to control the resistor network may also beimplemented as a non-periodic signal. For example, in one embodiment, adesired value of effective resistance may be obtained or achievedthrough proper choice of the average pulse width and average frequencyof the pulse waveform. Notably, one potential benefit of utilizing anon-periodic pulse waveform is that spurious content in its powerspectrum can be substantially reduced, which could be useful for certainapplications. In short, the signal to control the resistor network maybe a periodic signal and/or a pseudo-periodic or non-periodic signal(collectively referred to hereinafter as “non-periodic”), all of whichare intended to fall within the scope of the present inventions. Thus,although the discussions herein at times may illustrate or intimate aperiodic signal, the present inventions may be implemented usingnon-periodic signals to control the resistor network which are intendedto fall within the scope of the present invention; however for the sakeof brevity, such discussions, in the context of non-periodic signals maynot be repeated in detail.

In practice, the effective resistance of a switched resistor will alsobe impacted by parasitic capacitance, as indicated by FIG. 4C. A keyintuition here is that charge is transferred between the capacitances oneach side of the switch while it is on, and that charge then passesthrough the resistor on each side of the switch while it is off. Due tothis extra flow of charge, the average current is increased above theideal value such that the effective resistance is lowered. Due to thedistributed nature of capacitance on a practical resistor element, thereduction of effective resistance will nontrivially depend on T_(on) andT_(period) of the switch pulse signal. Although an analytical model cancertainly be derived, a straightforward method to estimate the actualeffective resistance is to simulate the switched resistor circuit (withthe inclusion of the estimated distributed capacitance as shown in FIG.4C) in transient operation with a circuit simulator such as SPICE. Inparticular, a voltage source can be placed across the terminals of thedevice, and the average current observed in simulation as a function ofT_(on) and T_(period).

When large effective resistance is desired, it may be advantageous toreduce parasitic capacitance through appropriate choice of the resistorelement (i.e., high resistance poly rather than low resistance poly), toreduce device parasitic capacitance by using the smallest switch size(i.e., smallest CMOS device) that still achieves a switch resistancethat is significantly less than the resistor element resistance, and tobe careful of leakage currents through the switch. Also, placing theswitch on one side of the resistor, as opposed to in the middle as shownin FIG. 4C, may lower the impact of parasitic capacitance. In general,reduction of T_(on) beyond a certain point will lead to diminishingreturns in increasing the effective resistance since the chargetransferred by parasitic capacitance will become a more significantportion of the average current in such embodiment. At that point,increase of T_(period) then becomes the primary tuning parameter forincreasing resistance further, wherein the effective resistance mayeventually become limited by leakage paths. For example, in 0.18 u CMOS,multiplication factors (i.e., T_(period)/T_(on)) of well over 20 areeasily achievable, such that greater than 10 MegaOhm effectiveresistance may be provided with 500 kOhm of poly resistance.

In one embodiment, the pulse width or pulse on-time (T_(on)) of theswitching signal is shorter than an RC time constant of the switchedresistor network. In this way, the effective resistance of the switchedresistor network increases as a function of one or more pulsingproperties (pulse width or frequency) of the switching or control signalof the switched resistor network.

Notably, it may be advantageous to employ circuitry, structures,architectures and/or techniques to increase the effective resistance ofa switched resistor. Indeed, all such circuitry, structures,architectures and/or techniques, whether now known or later developed,are intended to fall within the scope of the present inventions. Forexample, one such structure, architecture and/or technique is toincrease the number of switches utilized in the resistor network. (See,EXHIBIT 2 of the Provisional Application).

B. Exemplary Switched Resistor Loop Filter Topologies

Classical charge pump PLL topologies yield a loop filter transferfunction which is well understood and consists of the combination of alossy integrator (formed by the charge pump current integrating oncapacitor C₂) and a wider bandwidth feedforward path with gainR₁l_(pump) (and bandwidth w₂). An advantage of the high DC gain is thatit forces the average net current from the charge pump to be close tozero during steady state operation of the PLL, which leads to a wellcontrolled average phase error in the PLL phase detector. The primaryadvantages of such phase error control are that short pulses can beachieved at the charge pump output, which lowers the impact of referencespurs, thermal noise, and 1/f noise of the charge pump on the PLL outputphase noise. Due to mismatch between the current sources, the overallcharge pump characteristic will often be nonlinear, which introduces achallenge when trying to achieve low noise with fractional-N PLLstructures.

As shown in FIGS. 5A and 5B, loop filters implementing a switchedresistor network may achieve a similar transfer function as obtainedwith a charge pump loop filter, with the exception that the DC gain maybe constrained to be one. In the exemplary embodiment of FIG. 5A, thelow frequency pole, w₁, is primarily set by R₂ and C₂, the feedforwardgain is primarily set as the ratio C_(f)/(C_(f)+C₂), and the highfrequency pole, w₂, is primarily set by R₁ and C₁. In FIG. 5B, the lowfrequency pole is primarily set by R₃ and C₃, the feedforward gain isprimarily set by the ratio C_(f)/(C_(f)+C₃), and two high frequencypoles are primarily set by R₁, R₂, C₁, and C₂.

These two exemplary loop filters illustrate the versatility of theswitched resistor concept in addressing the needs of a variety of PLLapplications. One of the considerations seen in comparing the twodesigns is the need to consider switch resistance, which shouldgenerally be lower in value than the resistor value that is beingswitched. In the embodiment of FIG. 5A, the switches are placed withdirect connection to supply or ground, whereas in the embodiment of FIG.5B, the switches are also placed between resistor elements whose nodevoltages will typically fall in a mid-rail range (i.e., somewherebetween supply and ground). Notably, other supply and groundconfigurations are suitable.

Assuming that the switches are implemented with CMOS transistors, thelowest resistance of the switch is achieved when it directly connects tosupply or ground. As such, the switches in FIG. 5A can be implementedwith relatively small device sizes, and can be switched at fairly highfrequencies due to the small parasitic capacitance of the switches. Incontrast, the switches in FIG. 5B that are placed between resistorelements should be sized larger than the ones connected directly topower and ground, and the resulting parasitic capacitance limits howfast the switching can occur while still getting the desired boost ineffective resistance values. An advantage of the topology in FIG. 5B,however, may be that the out-of-phase switching of resistors R₁, R₂, andR₃ act to isolate the ripple caused by the pulsing action of the phasedetector such that excellent reference spur rejection is achieved. Inaddition to the requirement of larger switches, the cost of theout-of-phase switching technique is the introduction of delay in theloop filter. Notably, the additional delay may degrade phase margin inPLL applications demanding relatively high PLL bandwidth.

C. Exemplary Phase Detector Design

As discussed in the previous subsection, a significant differencebetween the loop filter provided by a charge pump PLL versus that of aswitched resistor PLL is that the DC gain can be quite large for thecharge pump embodiment, but may be constrained to be no more than onefor the switched resistor embodiment. In this subsection, we describeexemplary embodiments of how to adjust other components of the PLL toaccommodate lower DC gain which may be offered by the switched resistorloop filter. In particular, we propose exemplary high gain phasedetector (PD) implementations, and discuss the benefits of this approachin terms of noise performance of the PLL. Note that the VCO gain, Kv,may also be made larger to accommodate for those embodiments includinglower DC gain offered by the switched resistor loop filter.

To better understand the impact of DC gain of the loop filter, we firstexamine basic models of the charge pump and switched resistor PLLs asshown in FIGS. 6A and 6B. In the charge pump PLL, as shown in FIG. 6A,the loop filter transfer function, H(s), is modeled as the charge pumpcurrent, l_(pump), multiplied by the impedance of the loop filter RCnetwork, Z(s). The phase detector and divider are modeled as simple gainelements (note that gain can be less than one), and the VCO is modeledas an integrator with gain K_(v). In the switched resistor PLLembodiment, as shown in FIG. 6B, the modeling follows in a similarpattern to the charge pump PLL, with the subtle difference being thatthe loop filter transfer function, H(s), is simply defined as thevoltage gain through the RC network.

Due to the subtle difference in loop filter characterization, the PDgain may be computed in a slightly different manner for the twodifferent types of PLL structures. In each embodiment, an error signal,e(t), is formed at the output of the PD and averaged for each value ofthe phase difference,

_(error)(t). The difference between the PD gain calculation for the PLLstructures lies in how e(t) is formed. In the embodiment of the chargepump PLL, FIG. 6A indicates that e(t) corresponds to a pulse train thatalternates between −1, 1, and 0 depending on whether a down pulse, uppulse, or no pulses occur, respectively. In the embodiment of theswitched resistor PLL, FIG. 6B indicates that e(t) corresponds to theoutput of an RC network that is that is connected to ground, supply, orhas no connection to either depending on whether a down pulse, up pulse,or no pulses occur, respectively.

To provide a baseline understanding of how the PD gain will be increasedin the proposed PD designs, we first examine the PD gain of theclassical tristate PFD shown in FIG. 7. As mentioned earlier, the PDcharacteristic may be computed by sweeping the phase error acrossdifference values and then computing the average of the error signal,e(t), for each value. As shown in FIG. 7, the resulting PDcharacteristic is actually nonlinear in practice, but we can approximateit as linear under the assumption that the PLL is in lock and theinstantaneous phase variations due to noise and other perturbations arequite small. Therefore, in one embodiment, the PD gain is simplycalculated as the slope of the PD in the vicinity of zero phase error,which yields a value of 1/(2

) for the tristate PD.

FIG. 8 illustrates two proposed high gain PD structures that are wellsuited for the proposed switched resistor loop filters shown in FIGS. 5Aand 5B. The structure shown in FIG. 8A is well suited for the loopfilter in FIG. 5A due to the fact that its minimal circuit complexityallows a high reference frequency to be used, and the net delay throughthe structure is minimal so that a high PLL bandwidth can be supported.Note, in this embodiment, the delay shown in FIG. 8A need only be longenough to guarantee non-overlapping pulses in asserting the Up and Downsignals. In contrast, the structure shown in FIG. 8B is well suited forthe loop filter in FIG. 5B due to its support of multi-phase pulsing ofthe various switching signals in that loop filter. As stated earlier,the multi-phase switching technique reduces the voltage ripple caused bythe pulsing action of the phase detector before it influences thecontrol voltage of the VCO, thereby providing substantial reduction ofreference spurs.

A technique that is leveraged by the proposed high gain phase detectorsis to reduce the input range in phase error that is required to sweepacross the full output range of the PD output. As a baseline, considerthat the classical tristate PD shown in FIG. 7 requires an input phaserange of 4

to sweep across its full output range. In contrast, the PD shown in FIG.8A reduces this range according to the ratio of the pulse width of thedivider output to the period of the reference frequency. Therefore, theshorter the divider output pulses, the higher the PD gain that can beachieved. As for the PD shown in FIG. 8B, the input phase range isreduced according to the ratio of the nominal period of the divideroutput to the period of the reference frequency. In the embodiment ofthe PD shown in FIG. 8B, the divider frequency is set to a nominal valuethat is four times higher than the reference frequency, so that the PDgain is increased by a factor of 4 compared to the classical tristate PD(assuming Vdd=1 V).

As revealed by the above discussions, the proposed PD structures achievehigh gain by leveraging either (a) short pulses coming from the divideroutput (or reference output), or (b) a higher divider frequency than thereference frequency. In addition, one may want to the narrow the pulsesfurther in the case of signal “Last(t)” shown in FIG. 8B in order tofurther boost the effective resistance of R3 within the loop filtershown in FIG. 5B. To efficiently implement such PD structures, one alsoneeds to leverage appropriate techniques to create short pulses and/orleverage higher divider frequencies. We will address this issue in thenext subsection.

A subtle issue that occurs in implementing a PD with such a narrow inputphase range is that it can lead to longer or even dysfunctional behaviorwhen performing initial frequency acquisition for the PLL. As such, acomplementary frequency detection technique should be used inconjunction with the proposed high gain PD structures. We will discuss aproposed technique to achieve efficient frequency detection with thesestructures in a subsection to follow.

While exemplary phase detector embodiments are described and illustratedherein, it should be noted that other phase detector embodiments,structures, architectures and/or techniques, may be implemented inconjunction with the present inventions. Indeed, all phase detectorsconsistent with the present inventions, whether now known or laterdeveloped, are intended to fall within the scope thereof.

D. Considerations of Exemplary Divider Embodiments

In this section, we discuss various techniques related to the dividerdesign and short pulse generation. The first subsection describes aknown technique to generate short pulses from an asynchronous dividerstructure. The second subsection describes known techniques forsupporting divider output frequencies that are higher than the referencefrequency. Finally, the third subsection focuses on techniques forleveraging short pulse generation and sub-selection of pulses togenerate pulse waveforms for a switched resistor element.

Notably, while exemplary divider circuitry embodiments, structures,architectures and techniques are described and illustrated in detailherein, for example, divider circuitry providing a division factor of 4,the present inventions are not limited to such exemplary embodiments,structures, architectures and techniques and/or any particular divisionfactor. Indeed, all divider circuitry and/or techniques, whether nowknown or later developed (including, for example, division factors otherthan 4), may be implemented in conjunction with the present inventionsand, as such, are intended to fall within the scope thereof.

i) Narrow Width Pulse Generation Using the Divider

FIG. 9 illustrates a technique of generating short output pulses from acommonly used asynchronous divider circuit. As shown in the figure,pulses whose widths are 2^(n) multiples of the input period, where n=0,1, 2 . . . , can be readily tapped off from appropriate stages in thedivider using gate logic. In the example shown in the figure, tappingthe first divider stage yields output pulse widths of one input period,and tapping the second stage yields output pulse widths of two inputperiods. The advantage of this method of producing pulses is thatexcellent control is achieved of the pulse width since it is dependentto first order only on the input period, which is well controlled in thecontext of a PLL that is locked in steady-state. In contrast, pulsewidths created by delay circuits, as discussed below, are much moresensitive to temperature and fabrication process variations, as well assupply voltage changes.

FIG. 10 provides more detail on the Divide-by-2/3 stages that are usedin the asynchronous divider structure shown in FIG. 9. We see that thestages are composed of a small number of gates and latches. Thetransistor level implementation shown in FIG. 10 shows an efficientimplementation of the Divider-by-2/3 stages using CMOS devices.

ii) Using a Higher Frequency Divider Output

FIG. 11 shows the use of exemplary divider circuitry whose nominaloutput frequency is set to be four times that of the referencefrequency. When using the divider output in combination with the PDshown in FIG. 8B, the various PD pulses are produced according to therising edges of the divider and reference. Note that the details ofgenerating the “Pulse_Last” waveform will be discussed in the nextsubsection.

In examining FIG. 11, note that comparison of the output reference anddivider phases is performed through the Up and Down pulse generation,and that the remaining pulses of Pulse_Mid and Pulse_Last are used togate the charge through the loop filter so as to significantly reducereference spurs. As such, two out of every four divider edges are usedfor phase comparison, i.e., the edges adjoining the N₁ period shown inthe figure. The effective divide value seen by the PLL, N, correspondsto the sum of the four divide periods that occur per reference period,i.e., N=N₀+N₁+N₂+N₃.

FIG. 12 illustrates an exemplary method of controlling the exemplaryfour divide values to achieve a desired overall divide value of N. Inthis embodiment, the output of the divider is used to clock throughdifferent selection states of a digital multiplexer, which is fed theindividual divider values of N₀ through N₃. For a given desired overalldivide value, N, the integer portion is fed into digital mapping logicthat determines the values of N₀, N₁, and N₃, and also sets an offsetvalue N_(off). The fractional portion of N is send into a Sigma-Deltamodulator, which dithers its output such that its average corresponds tothe desired fractional value. This dithered divide value is added to thecomputed offset, N_(off), to form the value of N₂. Note that since theedge associated with the N₂ division value is not used to form the Up orDown pulses in the phase detector, the impact of Sigma-Delta ditheringon generating additional divider noise is reduced.

FIG. 13 shows one possible method to select each divide value N₀ throughN₃ by leveraging the multi-pulse PD shown in FIG. 8B. As shown in thefigure, a given divide value is selected when its respective enablepulse turns on. The advantage of this approach is that the divide valuescan be aligned as desired to their respective place within the referenceperiod. For instance, as mentioned earlier, it is desirable to placedivide value N₂, which is controlled by the Sigma-Delta modulator, awayfrom both of the divider edges that influence the Up and Down pulses.Note that since there is an inherent delay in the divide operation, theenable pulses gate through a given divide value in the previous dividerperiod.

iii) Additional Techniques for Pulse Generation

As described earlier in the text, the effective resistance value of aswitched resistor is a direct function of the ratio of the period ofswitching versus the on-time of the switch. As such, a switched resistorPLL will often need to employ techniques to appropriately set both ofthese parameters so as to achieve the desired effective resistance. Inthe subsection, we explore this issue in the context of controlling theswitching behavior of resistor R₃ in the example loop filter andmulti-pulse PD circuits shown in FIGS. 5B and 8B, respectively.

FIG. 14 illustrates the use of buffer delays to achieve desired pulsewidths and separation, along with frequency division to control theperiod of a switching signal for a given resistor. In the figure, bufferdelays 1, 2, and 3 are used to create separation between different pulsesignals in order to guarantee non-overlapping behavior between theon-times of those pulses. A non-overlapping condition between Up andDown pulses is desirable so as to avoid shoot-through currents in theirrespective switches, which has the negative impact of increasing powerconsumption and adding variability in the effective gain provided by thePD. A non-overlapping condition between the Up/Down, Mid, and Lastpulses is desirable so as to properly block ripple from previous loopfilter stages from impacting the loop filter output (i.e., VCO input) soas to minimize reference spurs. Note that adding explicit delay buffersto the Mid and Last signals, as shown in the figure, may be unnecessary,for example, if a non-overlapping condition between the various pulsesis met without such buffer delays.

In contrast to buffer delays 1 through 3, buffer delay 4 in FIG. 14 isused to control the width of the pulses in the Last signal, and adivide-by-N circuit is used to set the period of those pulses inincrements of the period of Up/Down pulses. For the example shown in thefigure, the Last pulses have much smaller width than that of the divideroutput, and period of the pulses is shown to be either the same (i.e.,N=1) or twice as long (i.e., N=2) as the period of the Up/Down pulses.

An alternative method of generating small pulse widths for the switchingsignals is to leverage the small pulses that can generated directly bythe divider output as discussed earlier. In the example shown in FIG.15, digital logic is leveraged to gate divider pulses to the Last signalat the appropriate phase (i.e., non-overlapping with the Mid signal) andperiod (i.e., integer multiples of the Up/Down period). Note that thepulses will, to first order, have the same pulse width in the Lastsignal as in the divider output. This technique has the advantage ofachieving well controlled pulse widths that are largely independent ofprocess and temperature variations since the divider output pulse widthsare set by the period of its input signal, which is well controlled whenthe PLL is in lock.

While exemplary techniques and circuitry for generating the signals forcontrolling the resistor network and/or switched resistor loop filterare described and illustrated herein, it should be noted that othercircuitry, architectures and/or techniques, may be implemented inconjunction with the present inventions. Moreover, while the control orswitching signals of several of the exemplary embodiments described andillustrated herein are generated by circuitry in the phase detectorcircuitry (see, for example, FIG. 33A), such signals may be generated bysignal generator circuitry not in the phase detector circuitry (see, forexample, FIG. 33B). Indeed, all circuitry, architectures and/ortechniques for generating signals to control the resistor network and/orswitched resistor loop filter, consistent with one or more aspects ofthe present inventions, whether now known or later developed, areintended to fall within the scope thereof.

Further, although exemplary circuitry, architectures and/or techniquesillustrates periodic signal for controlling the resistor network and/orswitched resistor loop filter, the control signal may be a non-periodicsignal. Indeed, in one embodiment, a non-periodic signal may provide aneffective resistance which is approximately achieved through properchoice of the average pulse width and average frequency of the pulsewaveform. Thus, the signal to control the resistor network and/orswitched resistor loop filter may be periodic, pseudo-periodic and/ornon-periodic, all of which are intended to fall within the scope of thepresent inventions. For the sake of brevity, implementations ofpseudo-periodic and/or non-periodic signals to control the resistornetwork and/or switched resistor loop filter are not discussedseparately in detail but such implementations are quite clear from thetext and illustration hereof.

E. Exemplary Frequency Detection Circuitry and Techniques

The present inventions may employ frequency detection circuitry. In thisregard, as with most PLL implementations, frequency detection circuitryprovides a lock condition for the PLL under starting conditions. Aswitched resistor PLL may employ a high gain PD. In one embodiment, wenow propose frequency detection circuitry and techniques that providerobust methods of obtaining initial lock in the PLL without impactingthe steady-state noise performance of the PLL.

FIG. 16 highlights issues of using a PD with increased gain whenperforming initial frequency acquisition in the PLL. The increased PDgain results in a reduced phase range over which the PD characteristicis linear. Since an offset in frequency between the reference anddivider output causes a ramp in phase error, initial frequencyacquisition will typically involve having the PD sweep through its PDcharacteristic at a rate that is in proportion to the amount offrequency error. A smaller linear phase range implies a faster effectivetraversal through the nonlinear regions of the PD for a given offset infrequency. For the given embodiment and assuming a relatively fasttraversal through the PD characteristic, the impact of R₃ will beminimal such that the effective gain of the loop filter appears to besmaller than one (note that the max gain of the loop filter is its DCvalue of one).

As an example, FIG. 16 shows an embodiment where the effective gain ofthe loop filter is given by the ratio C_(f)/(C_(f)+C₃), and we see thatthe relative perturbation of the voltage at the output of the filter ismuch smaller than the supply range. A concern of such a small voltageperturbation is that it may not include the required VCO voltage thatcorresponds to the desired output frequency, which directly compromisesthe goal of achieving frequency lock. In addition, the small linear PDrange exacerbates the nonlinear behavior of the PD as frequencyacquisition is occurring, which complicates the ability to achieve phaselock.

Although conventional frequency detection circuitry and techniques maybe suitable for a PLL including a resistor network and/or switchedresistor loop filter, in one embodiment of the present inventions, thefrequency acquisition circuitry and technique according to certainaspects of the present inventions provides relatively fast PLL lockingwithout compromising the noise performance of the PLL when it is inlock. As shown in FIG. 17, the frequency detector circuitry includesdigital logic that is used to compare edge counts of the reference anddivider output and a switched capacitor network that is used to adjustthe loop filter output based on the edge count comparison. In oneembodiment, we are assuming that the frequency divider has a nominaloutput frequency that is four times the reference frequency when the PLLis locked. Therefore, an error in frequency can be detected by checkingthat four divider edges occur per reference period. In the situationwhere less than four divider edges occur per reference period, theimplication is that the PLL output frequency is too low, whereas thesituation where more than four divider edges occurring per referenceperiod implies that the PLL output frequency is too high. A low or highPLL output frequency should be addressed by appropriately increasing ordecreasing the loop filter output voltage. The proposed frequencydetector performs this adjustment by charging a capacitor, C_(c), toeither the supply or ground and then connecting that capacitor to theloop filter capacitor at its output (i.e., C₃ in FIG. 17). As the loopfilter capacitor, C₃, connects to C_(c), its voltage is increasedaccording to the ratio C_(c)/(C_(c)+C₃) and the voltage on C₃immediately prior to the connection.

Notably, there are several advantages offered by the frequency detectorcircuitry embodiment of FIG. 17. For example, first, the implementationmay employ digital logic and a switched capacitor network. Second thereis no static current required in any of its elements, which facilitateslow power dissipation to be achieved. Third, the frequency detectorelements are disconnected from the loop filter unless a frequencydetection event occurs; in this way, the noise performance of the PLLwill not be impacted once the PLL is locked. Fourth, the frequencydetector may be always active, so if any unforeseen perturbation impactsthe PLL such that it loses lock, frequency acquisition will occur suchthat lock is re-obtained. Finally, it is worthwhile to note that themethod is not limited to having the nominal frequency correspond to fourtimes the reference frequency. Rather, the nominal divider frequency canbe any integer multiple, N, of the reference frequency, where is N>1,and the associated comparison count value may also be set to N. As anexample, if the divider frequency were designed to have a nominal valuethat is N=8 times the reference frequency, then the output of thecounter as shown in FIG. 17 may be compared to the value N=8 indetermining whether there is a frequency error.

While FIG. 17 provides a framework for the proposed frequency detectorimplementation, the process of counting edges need not be restricted toexplicit counter circuitry or structures. As an example, FIG. 18 showsan example of implementing the frequency detection logic by comparingthe edge locations of the reference as it propagates through cascadedregisters which are clocked by the divider output. Under lockconditions, edges of the reference should travel through four registersper reference period. By proper placement of the inputs of two XOR gatesto the register chain, we can sense if the reference edges traveledthrough three registers per reference period (i.e., output frequency istoo low) or five registers per reference period (i.e., output frequencyis too high). The output(s) of the XOR gates are fed into adjoininglogic to properly create the switched capacitor pulses, which should bedesigned to have non-overlapping characteristics.

FIG. 19 illustrates a system level simulation of frequency acquisitionof a switched resistor PLL using the proposed frequency detector in FIG.18. In this embodiment, the initial frequency of the PLL may be too low,so that several Charge_High(t) pulses (along with accompanyingConnect(t) pulses, which are not shown in the figure) may be employed tocharge up the loop filter output to the correct value. Once the PLLoutput frequency is close enough to its steady state value to avoidfurther cycle slipping, the frequency detector output pulsesautomatically stop and the phase detector becomes the sole means offeedback.

An advantage of combining the frequency detector with the multi-pulse PDas shown in FIG. 18 is that it may enhance immunity of the frequencydetector to falsely outputting frequency detection pulses to theswitched resistor network. To explain, when comparing edge countsbetween the reference and divider outputs, one should take care to haveproper phase alignment between the edges to avoid miscounts that couldbe falsely interpreted as frequency error. As shown at the top of FIG.20, good phase alignment places the reference and divider edgessufficiently far from each other in phase (or time) so that the edgecounting process yields instantaneous count values that match theaverage count value (i.e., four divider edges per reference edge in thisexample). As shown at the bottom of FIG. 20, bad phase alignmentcorresponds to the condition where jitter can alter the edge countingprocess so that the instantaneous count does not match the averagecount. In such case, false frequency detection signals will be generatedwhich will significantly degrade the noise performance of the PLL. Theadvantage offered by the frequency detector in FIG. 18 is that the phasedetector will insure proper phase alignment to avoid false frequencydetection events.

Proper phase alignment for edge counting may also be accomplished with asimple modification of the phase detector structure shown in FIG. 8A. Asshown in FIG. 21, if we design the divider such that its nominalfrequency is twice that of the reference frequency, than we can gate itspulses such that half are used for phase detection and half are used forfrequency detection (i.e., FD_Pulse in the figure). The benefit of theresulting frequency detection pulses are that they are placed relativelyfar away from the key reference edges, thus allowing a robust operationof counting edges.

In the PD implementation shown in FIG. 21, the frequency detectionpulses occur at the same frequency as the reference under lockconditions. However, although not necessary, it may be advantageous toutilize the same frequency detection logic as shown in FIG. 18 by simplydividing the reference frequency by four as shown in FIG. 22.

While exemplary inventive circuitry and techniques for phase detectionare described and illustrated herein, it should be noted that othercircuitry, architectures and/or techniques, may be implemented inconjunction with the present inventions—including, for example,conventional circuitry and techniques. Indeed, all circuitry,architectures and/or techniques consistent with one or more aspects ofthe present inventions, whether now known or later developed, areintended to fall within the scope thereof.

Moreover, while the switched capacitor frequency detection circuitry andtechniques have been described and illustrated in conjunction with theswitched resistor PLL circuitry and techniques of the presentinventions, the inventive switched capacitor frequency detectioncircuitry and techniques may also be implemented in conjunction withconventional type PLL circuitry, for example, circuitry employing acharge pump circuit. Here, the inventive switched capacitor frequencydetection circuitry and techniques facilitates rapid and robust lockcondition of the inventive PLL circuitry as well as conventional PLLcircuitry.

System Design Considerations of Exemplary Switched Resistor PLLEmbodiments

In this section, we describe certain system level consideration relatedto design of a switched resistor PLL. The first subsection describescertain considerations related to modeling of the PLL dynamics. Thesecond subsection describes modeling of noise in the switched resistorloop filter. The third subsection describes an issue of nonlinearitythat should be considered in design of a switched resistor PLL. Thefourth subsection describes a few practical issues such as enablingconfigurability in the switched resistor loop filter and the addition ofVCO test mode circuits.

A. Modeling of Steady-State PLL Dynamics

Modeling of the PLL dynamics of a switched resistor PLL may be similarto that of a classical analog PLL. In one exemplary embodiment, assumingthat the PLL is in lock, the phase detector may be modeled as a lineargain factor as previously discussed, and the presence of the frequencydetector can be ignored. As an example, FIG. 23 shows the linearizedmodel of a switched resistor PLL with a loop filter as shown in FIG. 5Band a multi-pulse phase detector as shown in FIG. 8B. As indicated inthe figure, the loop filter transfer function is composed of severalpoles and one zero, each of which can be calculated from the values ofcapacitance and effective resistance of the various loop filterelements. In certain embodiments, to achieve stability in the closedloop PLL response, the open loop gain of the PLL may be set such thatthere is adequate phase margin at the unity gain crossover of theoverall PLL open loop transfer function. In general, a reasonable phasemargin will be achieved when the unity gain crossover frequency occursbetween the zero, w₂, and pole, w_(p2), as indicated in FIG. 23.

B. Exemplary Noise analysis for Exemplary Switched Resistor PLL

Noise analysis of a switched resistor PLL may be performed in analogousfashion to that of a classical analog PLL. A distinguishingcharacteristic of a switched resistor PLL is the issue of how to modelthe impact of the switched resistor elements on the noise performance ofthe loop filter which implements the switched resistor network. As shownin FIG. 24, a first order approach to such analysis suggests the noiseof each switched resistor is simply approximated as a white noise sourcethat has variance 4kTR_(eff), where R_(eff) is the effective resistanceof the switched resistor element as discussed previously in this text.As such, note that the largest noise contributor in the loop filter willgenerally be the switched resistor element with highest effectiveresistance (i.e., R_(3_eff) for the example shown in FIG. 24). Anassumption in this noise modeling technique is that the parasiticcapacitance in each switched resistor element is negligible. If theparasitic capacitance is non-negligible, the switching operation willintroduce kT/C noise whose effect should be quantified by performingtransient noise simulations with a SPICE simulator.

Given the noise sources assumed for each switched resistor, a firstorder hand analysis for estimating the impact of loop filter noise onthe overall PLL noise performance can be performed by input referringthe loop filter noise sources to the input of the phase detector, andthen applying standard transfer function noise analysis on the overallPLL block diagram as shown in FIG. 25. For simplicity, the noiseanalysis can be focused on impact of the largest noise contributor(assumed to be the switched resistor corresponding to R_(3_eff) in thefigure).

Notably, to implement a more thorough noise analysis, it may bepreferable to construct a numerical model of the PLL in a program suchas Matlab so as to see the impact of noise across a broad range offrequencies. Also, the impact of VCO and reference noise may be readilyincluded in such an approach. An exemplary Matlab script performing thistask is provided in Appendix 1 of the Provisional Application.

C. Nonlinear Effects

A subtle issue related to implementing the switched resistor PLL withthe phase detector structures shown in FIG. 8 is the introduction ofnonlinearity in the phase comparison path. The cause of thisnonlinearity is generally attributed to the voltage dependent chargingcharacteristic of the switched RC network.

FIG. 26 illustrates the instantaneous voltage signal on the capacitor ofan RC network as the resistor is switched to ground and then the supplyvoltage with non-overlapping pulses. Defining T_(on) as the combinedwidth of the Up and Down pulses, and

as the amount of time that the Down pulse is increased and the Up pulseis decreased compared to the embodiment where they have equivalent pulsewidths, we can express the instantaneous voltage on capacitor C₁ atsample points following the Up/Down pulses as:

${V_{c\; 1}\lbrack k\rbrack} = {{{- V_{dd}}e^{{- \frac{1}{R_{1}C_{1}}}\frac{T_{div}}{2}}e^{\frac{1}{R_{1}C_{1}}\Delta\; T}} + {e^{- \frac{T_{div}}{R_{1}C_{1}}}{V_{c\; 1}\left\lbrack {k - 1} \right\rbrack}} + V_{dd}}$

Since

T directly varies with changes in phase error, the nonlinearrelationship between

T and the loop filter voltage, V_(c1)(t), also implies a nonlinearrelationship between phase error and the loop filter voltage. One shouldnote that the impact of the nonlinearity is significantly reduced as

T is made small relative to the RC time constant R₁C₁, as seen by:

$e^{\frac{1}{\;^{R_{1}C_{1}}}\Delta\; T} \approx {1 + \frac{\Delta\; T}{R_{1}C_{1}}}$

The issue of nonlinearity will not generally pose a problem forinteger-N phase-locked loops since changes in

T are primarily caused by noise so that

T<<R₁C₁ is valid. However, in certain instances, the nonlinearity shouldbe considered more carefully in the embodiment of fractional-Nphase-locked loops since the dithering of the divide value can causesignificant phase error perturbations. In general, the amount of phaseerror perturbation is a function of the PLL output frequency, dividerdesign, and choice of Sigma-Delta modulator since dithering of thedivider causes the phase to move in increments of the PLL output period.As indicated by the above equation, the switched resistor PLL topologiesshown in FIGS. 5 and 8 may have suitable nonlinearity where 1/(R₁C₁) ismade sufficiently small, though this will generally imply a lower PLLbandwidth.

FIG. 27 illustrates the impact of noise folding due to nonlinearity in aswitched resistor fractional-N PLL. Under ideal assumptions, thequantization noise due to divider dithering will be shaped to highfrequencies such that there is minimal quantization noise energy at lowfrequencies. The impact of passing the shaped quantization noise throughnonlinearity is to cause noise folding which generally increases theamount of quantization noise energy that occurs at low frequencies. Asshown in the example of FIG. 27, significant quantization noise canoccur at low frequencies due to the nonlinearity. However, since thereare other PLL noise sources, such as reference noise, divider jitter,VCO phase noise, and loop filter noise (from the resistive elements),the folding of quantization noise may remain below that of other noisesources (as shown in the figure). As discussed herein, lowering thebandwidth of the first RC stage in the loop filter (which implieslowering the PLL bandwidth), may reduce the impact of the nonlinearity.Also, lowering the quantization noise by decreasing the phase variationof the divider output (by increasing its input frequency or using anappropriately low order Sigma-Delta modulator topology) may also lowerthe impact of the nonlinearity (i.e., we want

T<<R₁C₁).

In some embodiments, an application will not allow the PLL bandwidthand/or divider phase variation to be lowered sufficiently to prevent theadverse impact of folded quantization noise due to nonlinearity. In suchembodiments, the proposed switched resistor PLL topology shown in FIG.28 would be worthy of consideration. In this structure, high linearityis achieved by using an XOR-based phase detector what has a singlebinary valued output, as opposed to Up and Down signals. In thisembodiment, the output of the PD can be fed directly into an RC filterthrough the use of a simple inverter. The first RC stage will be highlylinear and will act to filter out much of the high frequency energy dueto the shaped quantization noise from divider dithering. We can thenfollow the first RC stage with switched resistor network(s), where FIG.28 indicates the placement of a switched resistor at the second stage ofthe loop filter. The value of the switched resistor is to dramaticallyboost the resistance value so that area savings are achieved on theintegrated circuit. In this embodiment, the benefit of boosting thevalue of R₂ is that it helps in achieving a sufficiently low zero in thePLL transfer function such that closed loop stability of the PLL isassured.

Related to the issue of nonlinearity and quantization noise folding,there are a few additional considerations in designing a switchedresistor PLL. First, assuming that Up/Down pulses are leveraged as shownin FIG. 26, it may be advantageous to ensure that T_(on) does not changedynamically in time (i.e., due to divide value variations) since itwould otherwise introduce another source of nonlinearity in the phaseerror path. Prevention of such dynamic T_(on) variation can be achievedby maintaining a constant time difference between the key output divideredges, which is accomplished by making the associated divide valueconstant. For instance, for the divider control shown in FIG. 12, thevalue N₁ would be kept constant by choosing a different divide value(i.e., N₂ in the figure) to be controlled by the Sigma-Delta modulator.

The second consideration related to quantization noise folding is therelative frequency chosen for pulsing the switched resistors sincesubsampling shaped noise also leads to noise folding. As an example, forthe switched resistor loop filter shown in FIG. 28, the safest pulserate would be match the frequency of the phase error signal, e(t), sinceno subsampling of the quantization noise would occur. However, since thefirst RC filter stage acts to filter the quantization noise, it may beacceptable to allow subsampling if the impact on the overall PLL noiseperformance is acceptable. The advantage of subsampling (i.e., choosingthe period of the Pulse_R2(t) signal to be less than that of the phaseerror, e(t)) is that a larger effective resistance can be achieved forthe switched resistor, as discussed herein.

D. Configurability

In practice, it may be advantageous to have the capability to configurea given PLL implementation over a range of bandwidths and to allow testmodes to check PLL functionality. FIG. 29 shows one example of addingsuch features to a switched resistor PLL. In this embodiment, adjustmentof the PLL bandwidth is performed by allowing a change in thecapacitance C_(f) with the use of an array of switches and capacitors(C_(f_1) through C_(f_4)). Adjustment of C_(f) changes the open loopgain of the PLL, which changes the open loop unity gain crossover andthus the closed loop PLL bandwidth. In changing C_(f), one shouldgenerally change the stabilizing zero to maintain stability. In thisembodiment, we allow modification of the zero by providing an array ofswitched resistors (R_(3_1) through R_(3_4)) to realize R₃. Although notshown in the figure, we can also create an array of switched resistorsfor R₁ and R₂ to change the location of the higher frequency poles ofthe loop filter.

As for test modes, one useful feature is to be able to run the VCO inopen loop at different voltages in order to check its frequency and Kvvalue. The leftmost circuit in FIG. 29 allows different voltages to beplaced on the R₁ resistor. Assuming that we enable the switchesassociated with R₁, R₂, and R₃, and disable the Up and Down pulses aswell as frequency detection, we can then tune the VCO control voltageover a variety of settings to check its characteristics.

Another issue related to testing the VCO characteristics is to considerthe available range over which its tuning voltage can by varied whilethe PLL is in operation. As shown in FIG. 30, there are practicallimitations to the usable range of the phase detector, especially whendealing with a fractional-N phase-locked loop. For the example in thefigure, the usable range in the phase detector characteristic is abouthalf of the total theoretical range due to phase error variation fromthe Sigma-Delta modulation of the divider value. Since the DC gain ofthe loop filter may be equal to one for the switched resistor PLL, theusable varactor range may, in certain embodiments, be limited to half ofits theoretical range of the entire supply voltage. Note that classicalanalog phase-locked loops also have a varactor range, which is typicallyset by the charge pump rather than phase detector. In any case, testingof the VCO may benefit from knowledge of the achievable range bychoosing the control voltage values shown in FIG. 29 that coincide withthe limits in that range (i.e., ¼ and ¾ the voltage range in thisexample).

Design Considerations of Exemplary Switched Resistor PLLs

Key considerations of designing a switched resistor PLL may beidentification, selection and implementation of the size the loop filtercapacitances to achieve a predetermined PLL noise performance, and todesign the overall loop filter such that the PLL will be stable (forexample, over a predetermined range of operating frequencies). Otherissues such as achieving a VCO with a desired frequency range andsufficiently low phase noise and designing the high speed frequencydivider and Sigma-Delta modulator (in the embodiment of fractional-Nsystems, where N>0) for the proper PLL performance are also common to atraditional analog PLL design. One point of differentiation, however, isthat a switched resistor PLL will typically benefit from having a higherK_(v) of the VCO than what might be preferred in a traditional analogPLL due to the reduced DC gain of the switched resistor loop filter.

As indicated in FIG. 31, the noise of a switched resistor loop filterwill often be dominated by the largest resistor in its network, whichmay correspond to the key resistor that sets the stabilizing zero forthe PLL (i.e., R_(3_eff) in the figure). In order to reduce theR_(3_eff) noise, we need to reduce the value of R_(3_eff). To accomplishthis R_(3_eff) reduction, the value of C₃ should be increased since tofirst order, the stabilizing zero, w₂, corresponds to 1/(R_(3_eff)C₃).

In addition, as indicated in FIG. 31, the PLL unity gain crossoverfrequency (which corresponds to the closed loop PLL bandwidth) shouldtypically occur in the flat region of the loop filter transfer functionwhere the gain corresponds to C_(f)/(C_(f)+C₃) for the given example. Inthe embodiment where C₃ is increased in order to lower the R_(3_eff)noise, the value of C_(f) should also be increased in order to maintaina given unity gain crossover frequency (i.e., closed loop PLLbandwidth). While C_(f) will typically be smaller in value than C₃ inpractice, its implementation may be significant in chip area since itmay be beneficial to implement C_(f) with metal capacitors rather thanMOS gates in order to maintain a consistent capacitance value over avariety of bias conditions.

The tradeoff in noise performance versus capacitor area is common toboth switched resistor and traditional analog PLL implementations.Fortunately, the switched resistor PLL may have a much smaller capacitorarea requirement for a given level of noise due to the high gain offeredby its phase detector. As indicated in FIG. 31, the high PD gain reducesthe effective noise of the loop filter when referring it to the input ofthe PLL.

A detailed exemplary design procedure is provided in Appendix I of theProvisional Application in the form of a Matlab script that computes therequired resistor and capacitor values of the loop filter given otherPLL parameters. An example phase noise plot from this script is shown inFIG. 32, where the indicated phase noise performance is achieved withless than 100 pF of capacitance.

Importantly, the present inventions are neither limited to any singleaspect nor embodiment thereof, nor to any combinations and/orpermutations of such aspects and/or embodiments. For example, thepresent inventions are not limited to a switched resistor PLL asdescribed and illustrated above, but are also directed to the componentsthereof including, for example, the switched resistor network, switchedresistor loop filter circuitry, architectures, topologies andtechniques, phase detector circuitry and technique, divider circuitryand techniques, and/or circuitry, techniques and signals to control theswitched resistor network and/or switched resistor loop filtercircuitry. In this way, each of the aspects of the present inventions,and/or embodiments thereof, may be employed alone or in combination withone or more of the other aspects of the present inventions and/orembodiments thereof. For the sake of brevity, many of those permutationsand combinations are not discussed separately herein; however, allpermutations and combinations are intended to fall within the scope ofthe present inventions.

As such, the embodiments described and/or illustrated of the presentinventions are merely exemplary. They are not intended to be exhaustiveor to limit the inventions to the precise circuitry, techniques, and/orconfigurations disclosed. Many modifications and variations are possiblein light of the above teaching—including using certain conventionalcircuitry and techniques in conjunction with the inventive aspects of,for example, the switched resistor network, switched resistor loopfilter circuitry, architectures, topologies and techniques, phasedetector circuitry and technique, divider circuitry and techniques,and/or circuitry, techniques and signals to control the switchedresistor network and/or switched resistor loop filter circuitry. It isto be understood that other embodiments may be utilized and operationalchanges may be made without departing from the scope of the presentinventions. As such, the foregoing description of the exemplaryembodiments of the inventions has been presented for the purposes ofillustration and description. It is intended that the scope of theinventions not be limited solely to the description above.

For example, as noted above, while several of the exemplary embodimentsof the circuitry and techniques to generate control or switching signalsfor controlling the resistor network and/or switched resistor loopfilter have been described and/or illustrated herein as being integratedin the phase detector circuitry, the control or switching signals may begenerated by circuitry (here, a control signal generator) disposed in ornot disposed in the phase detector circuitry. (See, FIGS. 33A and 33B,respectively). All circuitry, architectures and/or techniques forgenerating signals to control the resistor network and/or switchedresistor loop filter, consistent with one or more aspects of the presentinventions, whether now known or later developed, are intended to fallwithin the scope thereof.

It should be noted that the term “circuit” may mean, among other things,a single component (for example, electrical/electronic and/ormicroelectromechanical) or a multiplicity of components (whether inintegrated circuit form, discrete form or otherwise), which are activeand/or passive, and which are coupled together to provide or perform adesired function. The term “circuitry” may mean, among other things, acircuit (whether integrated or otherwise), a group of such circuits, oneor more processors, one or more state machines, one or more processorsimplementing software, one or more gate arrays, programmable gate arraysand/or field programmable gate arrays, or a combination of one or morecircuits (whether integrated, discrete or otherwise), one or more statemachines, one or more processors, one or more processors implementingsoftware, one or more gate arrays, programmable gate arrays and/or fieldprogrammable gate arrays. The term “data” may mean, among other things,a current or voltage signal(s) whether in an analog or a digital form.

Moreover, “pulsing properties” of the switching or control signal toincrease the effective resistance of the switched resistor networkinclude the pulse width and frequency (or period) of such signal. Inaddition, the circuitry to generate the control or switching signalswhich changes (increases) the effective resistance of the switchedresistor network, whether or not disposed or integrated in the phasedetector circuitry, is collectively referred to in the claims ascircuitry of the phase detector circuitry (FIG. 33A), unless indicatedotherwise.

Notably, the various networks, circuitry described and/or illustratedherein (or portions and/or combinations thereof) may be integrated ormay be implemented using a plurality of discrete logic. All permutationsand/or combinations of integrated, discrete, hardwired and programmablecircuitry (which is programmed, for example, via software) forimplementing the circuits and circuitry are intended to fall within thescope of the present inventions. For example, in one embodiment, theswitched resistor network, switched resistor loop filter circuitry,architectures and topologies, phase detector circuitry, dividercircuitry, and/or circuitry to control the switched resistor networkand/or switched resistor loop filter circuitry may be integrated on amonolithic integrated circuit device. In other embodiments, one or morecomponents of the switched resistor PLL may be discrete or integrated ona monolithic integrated circuit device.

It should be further noted that the various circuits and circuitrydisclosed herein may be described using computer aided design tools andexpressed (or represented), as data and/or instructions embodied invarious computer-readable media, in terms of their behavioral, registertransfer, logic component, transistor, layout geometries, and/or othercharacteristics. Formats of files and other objects in which suchcircuit expressions may be implemented include, but are not limited to,formats supporting behavioral languages such as Matlab, Python, C,Verilog, and HDL, formats supporting register level descriptionlanguages like RTL, and formats supporting geometry descriptionlanguages such as GDSII, GDSIII, GDSIV, CIF, MEBES and any othersuitable formats and languages. Computer-readable media in which suchformatted data and/or instructions may be embodied include, but are notlimited to, non-volatile storage media in various forms (e.g., optical,magnetic or semiconductor storage media) and carrier waves that may beused to transfer such formatted data and/or instructions throughwireless, optical, or wired signaling media or any combination thereof.Examples of transfers of such formatted data and/or instructions bycarrier waves include, but are not limited to, transfers (uploads,downloads, e-mail, etc.) over the Internet and/or other computernetworks via one or more data transfer protocols (e.g., HTTP, FTP, SMTP,etc.).

Indeed, when received within a computer system via one or morecomputer-readable media, such data and/or instruction-based expressionsof the above described circuits may be processed by a processing entity(e.g., one or more processors) within the computer system in conjunctionwith execution of one or more other computer programs including, withoutlimitation, net-list generation programs, place and route programs andthe like, to generate a representation or image of a physicalmanifestation of such circuits. Such representation or image maythereafter be used in device fabrication, for example, by enablinggeneration of one or more masks that are used to form various componentsof the circuits in a device fabrication process.

Moreover, the various circuits and circuitry, as well as techniques,disclosed herein may be represented via simulations using computer aideddesign and/or testing tools. The simulation of the switched resistorPLL, and/or components thereof, for example, the switched resistornetwork, switched resistor loop filter circuitry, architectures andtopologies, phase detector circuitry, divider circuitry, and/orcircuitry to control the switched resistor network and/or switchedresistor loop filter circuitry (or portions of the foregoing), and/ortechniques implemented thereby, may be implemented by a computer systemwherein characteristics and operations of such circuitry, and techniquesimplemented thereby, are imitated, replicated and/or predicted via acomputer system. The present inventions are also directed to suchsimulations of the switched resistor PLL, and/or components thereof,and/or techniques implemented thereby, and, as such, are intended tofall within the scope of the present inventions. The computer-readablemedia corresponding to such simulations and/or testing tools are alsointended to fall within the scope of the present inventions.

Notably, for the avoidance of doubt, the present inventions are alsoapplicable to delay-locked loops or other clock alignment circuitry thatimplements one or more of the switched resistor network, switchedresistor loop filter circuitry, architectures and topologies, phasedetector circuitry, divider circuitry, and/or circuitry to control theswitched resistor network and/or switched resistor loop filter circuitrydescribed and/or illustrated herein. It is intended that suchdelay-locked loops or other clock alignment circuitry fall within thescope of the present inventions.

What is claimed is:
 1. A phase locked loop circuit comprising:comparison circuitry to receive a reference signal and an oscillationsignal and to generate an error signal according to a relationshipbetween the reference signal and the oscillation signal; a loop filterto receive an output of the comparison circuitry, the loop filter havingan impedance network that is switched as a function of the error signalso as to vary impedance properties of the impedance network; andoscillation circuitry to receive an output of the impedance network andto responsively generate an output signal; wherein the oscillationsignal is a function of the output signal.
 2. The phase locked loopcircuit of claim 1, wherein the comparison circuitry comprises a phasedetector that is to produce an up/down signal as a function of phaseerror between the reference signal and the oscillation signal, andwherein the oscillation circuitry comprises a voltage controlledoscillator that is driven according to an accumulation of integratedphase error between the oscillation signal and the reference signal. 3.The phase locked loop circuit of claim 2, wherein the impedance networkcomprises at least one metal oxide semiconductor (MOS) device to beswitched according to pulses of a control signal driven as a function ofchange in sign of the up/down signal.
 4. The phase locked loop circuitof claim 3, wherein the impedance network comprises at least onepolysilicon resistor coupled in series with the MOS device, an effectiveresistance of the impedance network being dependent on the at least onepolysilicon resistor.
 5. The phase locked loop circuit of claim 1,wherein the comparison circuitry comprises a phase frequency detectorthat is to drive an accumulator to adjust control of the oscillationcircuitry at separate adjustment rates for each of phase difference andfrequency difference between the reference signal and the oscillationsignal.
 6. The phase locked loop circuit of claim 1, wherein thecomparison circuitry comprises a digital phase detector circuit.
 7. Thephase locked loop circuit of claim 1, wherein the impedance networkcomprises at least one resistor, at least one capacitor and at least oneswitch, and wherein the switch is to be switched at a rate such that theimpedance network reduces 1/f noise of the oscillation circuitry, wheref is a frequency of oscillation of at least one of the reference signaland the oscillation signal.
 8. The phase locked loop circuit of claim 1,wherein the phase locked loop circuit further comprisesfrequency-divider circuitry to divide the output signal by a factor ofat least four, to generate the oscillation signal.
 9. The phase lockedloop circuit of claim 1, wherein the impedance network is a resistancenetwork.
 10. The phase locked loop circuit of claim 1, wherein: thephase locked loop circuit further comprises at least one adjustablecapacitor and circuitry to selectively adjust an effective capacitanceprovided by the at least one adjustable capacitor; and a loop bandwidthof the phase locked loop circuit is dependent on the effectivecapacitance.
 11. The phase locked loop circuit of claim 1, wherein theimpedance network is to be switched according to a pulsed control signalhaving a pulse duration less than an RC time constant of the impedancenetwork.
 12. An integrated circuit, comprising: a chip; a phase lockedloop circuit on the chip; the phase locked loop circuit comprisingcomparison circuitry to receive a reference signal and an oscillationsignal and to generate an error signal according to a relationshipbetween the reference signal and the oscillation signal, a loop filterto receive an output of the comparison circuitry, the loop filter havingan impedance network that is switched as a function of the error signalso as to vary impedance properties of the impedance network, andoscillation circuitry to receive an output of the impedance network andto responsively generate an output signal; wherein the oscillationsignal is a function of the output signal.
 13. The integrated circuit ofclaim 12, wherein the comparison circuitry comprises a phase detectorthat is to produce an up/down signal as a function of phase errorbetween the reference signal and the oscillation signal, and wherein theoscillation circuitry comprises a voltage controlled oscillator that isdriven according to an accumulation of integrated phase error betweenthe oscillation signal and the reference signal.
 14. The integratedcircuit of claim 13, wherein the impedance network comprises at leastone metal oxide semiconductor (MOS) device to be switched according topulses of a control signal driven as a function of change in sign of theup/down signal.
 15. The integrated circuit of claim 14, wherein theimpedance network comprises at least one polysilicon resistor coupled inseries with the MOS device, an effective resistance of the impedancenetwork dependent on the at least one polysilicon resistor.
 16. Theintegrated circuit of claim 12, wherein the comparison circuitrycomprises a phase frequency detector that is to drive an accumulator toadjust control of the oscillation circuitry at separate adjustment ratesfor each of phase difference and frequency difference between thereference signal and the oscillation signal.
 17. The integrated circuitof claim 12, wherein the comparison circuitry comprises a digital phasedetector circuit.
 18. The integrated circuit of claim 12, wherein theimpedance network comprises at least one resistor, at least onecapacitor and at least one switch, and wherein the switch is to beswitched at a rate such that the impedance network reduces 1/f noise ofthe oscillation circuitry, where f is a frequency of oscillation of atleast one of the reference signal and the oscillation signal.
 19. Theintegrated circuit of claim 12, wherein: the phase locked loop circuitfurther comprises at least one adjustable capacitor and circuitry toselectively adjust an effective capacitance provided by the at least oneadjustable capacitor; and a loop bandwidth of the phase locked loopcircuit is dependent on the effective capacitance.
 20. The integratedcircuit of claim 12, wherein the impedance network is to be switchedaccording to a pulsed control signal having a pulse duration less thanan RC time constant of the impedance network.
 21. A method of operationin a phase locked loop circuit having comparison circuitry, a loopfilter having an impedance network, and oscillation circuitry, themethod comprising: comparing a reference signal and an oscillationsignal with the comparison circuitry and responsively generating anerror signal according to a relationship between the reference signaland the oscillation signal; receiving an output of the comparisoncircuitry with the loop filter and using an output of the loop filter tocontrol the oscillation circuitry; and switching the impedance networkas a function of the error signal so as to vary impedance properties ofthe impedance network and thereby adjust the output of the loop filter;wherein the oscillation signal is a function of the output signal.